In recent years, software-defined radios that use general-purpose hardware and that can switch between radio communication standards that only correspond to settings on software have been actively studied and developed. Software-defined radios need to deal with radio frequencies from several 10 MHz to several GHz that are generally used.
FIG. 1 shows the structure of a receiver disclosed in Non-Patent Literature 1 as an example of a receiver that receives RF (Radio Frequency) signals. In this receiver, a received RF signal is input through an antenna to an RF circuit composed of band pass filter 280, low noise amplifier (LNA) 281, RF tracking filter 282, and frequency converter 283. Band pass filter 280 eliminates interference signals that lie in an unnecessary bandwidth from the received RF signal so as to prevent the downstream circuits from getting saturated (however, in this case, band pass filter 280 cannot eliminate interference signals having frequencies that are close to the frequency of the desired signal). The received RF signal that passes through band pass filter 280 is amplified by LNA 281. After RF tracking filter 282 further suppresses the remaining interference signals, frequency converter 283 converts the frequency of the received RF signal using a clock signal generated by clock generator 284, and then a baseband section performs signal processes such as filtering for the resultant signal.
From the point of cost and the size of circuit area for software-defined radios, it is not preferred that components that differ in characteristics be implemented and switched between applicable radio communication standards. In particular, reducing the number of band pass filters that are integrated in a chip is difficult and has become a critical technical issue so as to accomplish software-defined radios. To reduce the number of band pass filters, a technique that allows the pass bandwidth of a band pass filter to become variable or another technique that allows signals having frequencies of several 10 MHz to several GHz to pass may be considered. On the other hand, band pass filters located upstream of the LNA need to satisfy both high linearity and low noise characteristics. Although passive filters such as surface acoustic filters (SAWs) excellently satisfy such characteristics, it is difficult to adjust the pass bandwidth of passive filters.
Thus, in receivers applicable for software-defined radios, SAW filters that have wide pass bandwidths might be a hopeful candidate for band pass filters. However, in this case, depending on the frequency of a desired signal, interference signals having frequencies up to 10 times higher than the frequency of the desired signal could be input to the LNA and the frequency converter. Thus, an RF circuit needs to have high linearity that can stand very strong interference signals. While CMOS process has been miniaturized, power supply voltage has been decreased, and the dynamic range of the RF circuit has been narrowed, accomplishing high linearity has become a very important technical issue.
As a technical issue for frequency converters, interference signals having frequencies that are close to high order harmonics of an LO signal are frequency-converted into baseband signals along with the desired signal because of harmonics of the LO signal and nonlinearity of a mixer. Such interference signals cannot be eliminated by ordinary frequency converters that have dull frequency characteristics and thereby they can narrow the dynamic range. In particular, if the LO frequency of the LO signal is low, it is difficult to transmit the LO signal as a sine wave. Rather, it would be advantageous to transmit the LO signal as a square wave from a point of reduced the size of circuit area and reduced power consumption. However, since an LO signal having a square waveform contains many odd-order harmonics, the lower the LO frequency, the more the foregoing issue becomes serious. On the other hand, although a receiver that has a differential structure can eliminate interference signals having frequencies of even-order harmonics of the LO signal, if the differential structure is asymmetrical, it would become difficult to sufficiently suppress interference signals. As a result, in this situation, the dynamic range would be narrowed.
The receiver disclosed in Non-Patent Literature 1 uses both a mixer (FIG. 2) called harmonics eliminating mixer located in frequency converter 283 and RF tracking filter 282 so as to solve the foregoing problem. The harmonics eliminating mixer uses a three-phase square LO signal having phases that vary by 45 degrees each. For example, a base band signal having a phase of 0 degree (hereinafter referred to as base band I signal) is obtained by multiplying the received RF signal by an LO signal having phases of −45 degrees, 0 degree, and 45 degrees, weighting the results with gains of 1, √2, and 1, respectively, and adding the results. A base band signal having a phase of 90 degrees (hereinafter referred to as base band Q signal) can be obtained by multiplying the received RF signal by an LO signal having phases of 45 degrees, 90 degrees, and 135 degrees, respectively, weighting them with the foregoing gains, and adding the results. Likewise, inverted signals of base band I signal and base band Q signal can be obtained by using an LO signal having phases of 135 degrees, 180 degrees, and 225 degrees and an LO signal having phases of 225 degrees, 270 degrees, and 315 degrees, respectively. In other words, to demodulate base band I signal and base band Q signal, an LO signal having a total of eight phases that vary by 45 degrees each is used. By weighting an LO signal having phases that vary by 45 degrees each and adding the results, the frequency conversion gain of frequency converter 283 becomes (1+√2·z−1+z−2) where z−1 is a time delay corresponding to a phase of 45 degrees of the LO signal. Thus, frequency converter 283 has a finite impulse response (FIR) filter characteristic of three taps (1+√2·z−1+z−2). The standardized frequency of the FIR filter is eight times higher than the LO frequency. The FIR filter has zero gain points at frequencies that are three times and five times higher than the LO frequency. As a result, the FIR filter can eliminate interference signals having frequencies that are three times and five times higher than the LO frequency (FIG. 3).
As another example, FIG. 4 shows a frequency converter (Patent Literature 1). The frequency converter has a generalized harmonics eliminating mixer that can eliminate higher order harmonics than the harmonics eliminating mixer shown in FIG. 2. The harmonics eliminating mixer disclosed in Patent Literature 1 is driven with discrete LO signals that are shifted by different angles from a common LO signal. The harmonics eliminating mixer has (2w-1−1) discrete mixers ((2n+1) discrete mixers in FIG. 4) such that output signals of the individual discrete mixers are added and output. Since the gain in which output signals of the individual mixers are weighted and added is proportional to cosine values corresponding to the phases of the individual LO signals, interference signals having frequencies of odd-order harmonics, up to (2w−3)-th order harmonics, of the LO signal can be eliminated.